With the ever-increasing need for hyper-rate communication systems and with the expanding utility of personal computers, more convenient and less expensive communication systems, with high data rates, are required. Such a need has been met through digital subscriber line communication systems, abbreviated as “xDSL”, using conventional copper telephone lines commonly present in homes and offices.
xDSL refers generally to all forms of communication that employ telephone lines, for example, high data-rate DSLs (HDSL) that serve as a substitute for conventional T1 lines, symmetric DSLs that replace T1 or E1 lines by one twisted pair copper line, and asymmetric DSLs capable of transferring a large amount of data in the environment of public switched telephone networks (PSTN).
In ADSL system, the term “asymmetric” arises from the fact that downstream data to be transferred in the direction from a central station CO to a remote terminal has a wider bandwidth and a larger data size as compared to upstream data to be transferred in the direction from the remote terminal to the central station. In the ADSL system, data communication and plain old telephone service (POTS) can be transferred at the same time. In contemporary systems, normal data rates of the ADSL system are 8 Mbps in the downstream direction and 640 Mbps in the upstream direction.
Unfortunately, modulated signals emit energy to adjacent copper wires within the same cable bundle when electrical energy is transmitted through copper wires in a telephone network. This phenomenon of electromagnetic cross-coupling is referred to as “crosstalk”. In a typical telephone network, a pair of copper wires insulated from other are joined by a cable binder. At certain transfer rates, pulsating crosstalk interference occurs at a non-negligible level between adjacent systems in cable bundles transceiving information within the same frequency range, resulting in distortion of signal waveforms.
Crosstalk may be classified into two categories. Near-end crosstalk, referred to as NEXT, is an important factor because it causes remarkably large crosstalk on the original signal as a result of a high energy signal being propagated through a nearby system. Far-end crosstalk, referred to as FEXT, is evaluated by measuring disturbance at a far end of a transmission medium after transferring a test signal through a pair of copper wires from an end of a channel. FEXT is therefore useful for determining crosstalk noise for data communication with devices located at counter ends of the copper wires. The dimension of the FEXT is generally smaller than the NEXT because an interference signal of the FEXT degrades during transport along the copper wires during signal transmission.
ADSL modems come in many forms, including annex-A, annex-B, and annex-C in accordance with application environments. The annex-A modem includes the steps of HS/T1413, training, channel analysis, messaging, exchanging, and showtime. A phase-locked loop (PLL) operation is carried out in the training step to set loop timing between a central station and a terminal, and should be continuous to retain the loop timing. The annex-A modem uses the 64'th one among 256 carriers for the PLL operation, i.e., tone No. 64 (hereinafter, referred to as #64 tone), and encodes channels in the form of quadrature amplitude modulation (QAM) up to 15 bits at maximum. A transceiving signal during an initialization process, before HS or T1413, is provided according to a 4QAM procedure using two bits. After completing the process of HS or T1413, the central station sends constellation information of (+1, +i) with the #64 tone. The constellation information on the #64 tone is successively retained without disconnection but during an echo cancel training period. A remote terminal may restore the same timing with the central station by means of the constellation information about the #64 tone.
An ADSL receiver of a remote terminal 10, shown in FIG. 1, includes an analog-to-digital converter (ADC) 11, a time-domain equalizer (TEQ) 12, a serial-to-parallel (S/P) converter 13, a fast Fourier transformer (FFT) 14, a frequency-domain equalizer (FEQ) 15, a QAM decoder 16, a digital phase lock loop (DPLL) 17, a digital-to-analog (DAC) converter 18, and a voltage-controlled crystal oscillator (VCXO) 19. The DPLL 17 includes a phase detector 21 and a loop filter 22. The DPLL 17, the DAC 18 and the VCXO 19 form a clock recovery loop in the ADSL receiver.
The ADSL system generally uses a discrete multi-tone (DMT) coding technique associated with providing a multiplicity of channels for information transmission. The DMT offers, for example, 256 independent sub-channels (or tones) divisionally assigned to the bandwidth from 0 kHz to 1.104 MHz at intervals of 4.3125 kHz. The bandwidth of 0˜20 kHz is used for the plain old telephone service (POTS) region.
As shown in FIG. 1, an analog signal RX arriving through a data transmission channel (e.g., a telephone network) is applied to the ADC 11. The ADC 11 converts the received analog signal RX into a digital signal. The converted digital signal is applied to the TEQ 12.
The TEQ 12 removes portions of inter-symbol interference (ISI) of the digital signal generated by the ADC 11. A data stream synchronized in a predetermined time domain is applied to the S/P converter 13 from the TEQ 12. The S/P converter 13 receives and stores the serial data stream in sequence and outputs the stored samples in parallel by N packets (e.g., 256 packets). The 256 samples are provided to 256-point FFT 14 to be converted into frequency-domain symbols. The frequency-domain symbols are applied to the FEQ 15. The FEQ 15 corrects amplitudes and phases of the symbols and the corrected symbols are applied to the QAM decoder 16. The QAM decoder 16 carries out a QAM decoding operation against the input symbols and outputs reception data RD.
In order to maintain the continuity between the DMT symbols, the frequency of a sampling clock signal of the ADC 11 in the remote terminal 10 should be integer times of a pilot tone (i.e., #64 tone) frequency. For instance, when the frequency of the pilot tone frequency output from the FEQ 15 is 276 kHz and the sampling frequency is 2.208 MHz, one period of the pilot tone is composed of 8 samples. Therefore, the sampling clock signal of the ADC 11 can be obtained by abstracting an eight times frequency from the pilot tone frequency which has been synchronized to a reference signal REF.
The phase detector 21 in the DPLL 17 compares the reference signal REF with the pilot tone that the #64 tone provided from the FEQ 15. Here, the frequency of the reference signal REF is 276 kHz and the constellation value for the pilot tone is (+1, +i). When the constellation value of the reference signal REF is (Xref, Yref) and the constellation value of the pilot tone output from the FEQ 15 is (Xr, Yr), a phase error PE is defined as follows.PE=tan−1≈Xr−Yr  Equation 1
The phase detector 21 generates a signal corresponding to a phase difference between the pilot tone z64 and the reference signal REF. The loop filter 22 is formed of a secondary-order active loop filter, preferably designed with a parameter value established by considering acquisition times and tracking errors.
The DAC 18 converts a digital signal of the DPLL 17 into an analog signal. The VCXO 19 generates a sampling clock to the ADC 11 in response to the analog signal provided by the DAC 18.
FIG. 2 enumerates various types of transceiving signals defined by the ITU-T G.992.1 standard. In the chart of FIG. 2, ATU-C represents signals transmitted to a remote terminal from a central station while ATU-R represents signals transmitted to a central station from a remote terminal.
The G.992.1 annex-A standard defines signals that conduct training processes for modules constructed in a transceiver and an initialization process after the training processes. In FIG. 2, the pilot signal represents the 4QAM signal of 276 kHz mapped at constellation (+1, +i) and the REVERB signal is a pseudo-random signal having a phase difference of 180 degrees from the pilot signal. After completing a clock recovery operation between the central station and the remote terminal by means of the pilot signal, the training process of the FEQ 15 is carried out by means of the REVERB signal. A procedure of the training is segmented into multiple states in accordance with a signal to be transmitted, in which timing recovery should precede the equalizer training procedure. A signal generated by the FFT 14 after being sampled by a fixed clock signal, Y, is represented as follows.Y=H×X  Equation 2
In the Equation 2, the parameter X denotes a signal received from the central station, i.e., the REVERB signal, and the parameter H denotes a channel response. The ITU-T G.992.1 standard defines the REVERB signal as a periodic signal. Thus, the channel response H can be obtained from Y/X. It is generally known to apply the inverse of H to the equalizer and a correct channel response can result from securing a timing recovery.
As aforementioned, it is inevitable for ADSL network lines to be affected by crosstalk when they coexist with other communication lines within the same bundle of cable. FIG. 3 shows various patterns of crosstalk as an example, by which it is difficult to set an exact timing recovery. Moreover, as shown in FIG. 4, such crosstalk causes phase differences between the pilot tones on center of 45 degree that corresponds to the constellation (+1, +i).
FIG. 4 shows phase profiles of the pilot tones received through ADSL network lines, comparatively plotting them with crosstalk and without crosstalks, which are results of the PLL operation using the #64 tone among the reception signal (e.g., C-REVERB2 in FIG. 2) accepted during the training process. As shown in FIG. 4, the phase of the #64 tone is close to: 45 degrees when there is no crosstalk, while the phase considerably strays from 45 degrees when there is crosstalk. Such results arise from physical characteristics of telephone lines in which the SNR of the #64 tone; is reduced by crosstalk effects.
As described above, since the conventional ADSL receiver is designed to carry out the PLL operation using the #64 tone, deterioration of SNR of the #64 tone may cause fluctuations in other channels. As a result, error rate after completing the equalization by the FEQ 15 is affected by the sum of phase transitions due to loop timing errors and physical channel noises. In this case, it is difficult to evaluate SNRs of sub-channels because of the physical characteristics of channels and because the SNRs of sub-channels are substantially degraded due to the loop timing errors.
In addition, as a data rate of system is proportional to the SNR, a lower SNR may cause the data rate to be insufficient to meet with its desired level. Therefore, minimization of the deterioration of SNR to enhance the data rate permissible in physical conditions of the channel is desired.